1. Field of the Invention
The present invention relates to radio communication systems, and more particularly to a radio receiver, and a method therefore, for efficiently suppressing attenuation properties of a low frequency signal.
2. Description of the Related Art
Generally, in an analog frequency modulation type radio communication systems such as, for example, an analog cellular phone or an analog radio telephone, the typical frequency demodulation mode used in a radio receiver is a heterodyne mode employing a frequency discriminator. A heterodyne receiver is a receiver which heterodynes mixes a received high frequency signal with a local oscillator signal in the receiver and obtains an output signal having the frequency of the difference and the sum between frequencies of both signals to perform amplification and detection of the output signal.
FIG. 1 is a block diagram showing the construction of a typical heterodyne receiver.
Referring to FIG. 1, a radio frequency, electromagnetic signal, i.e., a high frequency signal having a spectrum of a double side band is received by antenna 101 which converts the high frequency signal into an electrical signal which is supplied to radio frequency filter 103. Radio frequency filter 103 is a wide band filter which generates a filtered signal. The filtered signal is comprised of signal portions of the electrical signal applied thereto which have frequencies within the passband of radio frequency filter 103. The filtered signal generated from radio frequency filter 103 is applied to first down-mixer 105. Mixer 105 also receives an oscillating signal generated by first local oscillator 152. First down-mixer 105 mixes the filtered signal and the local oscillator signal to generate a down-mixed signal, i.e., the first intermediate frequency signal(or "1st IF) which is applied to bandpass filter 107. Bandpass filter 107 generates a filtered signal which is applied to preamplifier 109 which amplifies the filtered signal. The amplified signal from preamplifier 109 is applied to second down-mixer 160. Mixer 160 also receives an oscillating signal generated by second local oscillator 146. Second down-mixer 160 mixes the amplified signal and the oscillating signal to generate a down-mixed signal, i.e., the second intermediate frequency signal(or "2nd IF) which is applied to bandpass filter 161. since both the first and the second intermediate frequency signals are not 0 Hz frequencies, such a receiver is called a heterodyne receiver. Bandpass filter 161 allows the frequency band given for an allocated channel of the second intermediate frequency signal to be filtered, and limiter 162 amplifies the amplitude of the filtered, second intermediate frequency signal to the amplitude suitable for demodulation. Frequency demodulator 131 generates a demodulated signal which is supplied to audio processor 133. Audio processor 133 allows an original aural signal to be outputted through a transducer such as speaker 135 by performing functions, such as, for example, audio filtering, de-emphasis, and variable gain for the demodulated signal.
Referring to only the drawing, the heterodyne receiver shown in FIG. 1 is likely to be seen as a very simple construction. However, it is to be noted that bandpass filter 161 as a channel filter is realized with a ceramic or LC filter occupying a relatively large space. Thus, since such a bandpass filter 161 is essentially not integrated by a semiconductor, it is difficult to reduce the volume of the receiver from a manufacturing standpoint. In addition, another problem with the prior art is that a heterodyne receiver employs an asynchronous system and as such, receiving sensitivity as a measurement for performance of the receiver is relatively degraded.
Meanwhile, in order to resolve these conventional problems, the Barber receiver as shown in FIG. 2 has been designed. This type of receiver, which was proposed by Barber in 1947, is seen about 1.5 times as complex as the receiver shown in FIG. 1. But, substantially, since it is possible to apply modern semiconductor integration technology to the Barber receiver, it is more attractive than the heterodyne receiver.
Referring now to FIG. 2, the procedure in which a high frequency signal having a spectrum (301,302 in FIG. 3a) of a double sided band is received by antenna 101 and the first intermediate frequency signal is generated before preamplifier 109 is similar to that the heterodyne receiver of FIG. 1. However, with respect to the subsequent procedure in which the second intermediate frequency signal is generated, the operation of the Barber receiver is very different from that of the heterodyne receiver. The first intermediate frequency signal 110 outputted from preamplifier 109 is applied to second down-mixer 111 of an I-channel and second down-mixer 119 of a Q-channel with the electric power being equally divided into halves. Such a receiver differs from the heterodyne receiver in that it complex-converts the intermediate frequency while dropping the intermediate frequency to 0 Hz, divides a channel into two mutual orthogonal vector channel(I-channel, Q-channel) to channel-filter the frequency at a baseband and demodulate the frequency. The Barber receiver is commonly referred to as a zero IF(i.e., an intermediate frequency of 0 Hz) receiver.
The output signal from second local oscillator 146 is applied to 90.degree. phase shifter 147, which applies the output signal to second down-mixer 111 of the I-channel with a phase delay of 0.degree. and second down-mixer 119 of Q-channel with a phase delay of 90.degree., respectively. At this moment, the oscillating frequency from second local oscillator 146 is equal with the frequency of the first intermediate frequency signal. Therefore, second down-mixer 111 generates a baseband signal with a phase delay of 0.degree. on line 112, and second down-mixer 119 generates a baseband signal with a phase delay of 90.degree. on line 120. Lines 112 and 120 are coupled to filters 113 and 121, respectively.
Immediately prior to its division, the first intermediate frequency signal on line 110 maintains a double sided band as it is. This is because only the centered frequency of the first intermediate frequency signal has been down-converted to a frequency besides 0 Hz from the centered frequency of the high frequency signal having a spectrum of a double sided band received by antenna 101. However, since second down-mixers 111 and 119 down-converts the center frequency to 0 Hz, both lower side band and upper side band of a spectrum of the first intermediate frequency signal are representative of two single side bands having band limiting properties of a distance spaced apart equally from the center frequency. The lower side band is of a negative spectrum and the upper side band of a positive spectrum (306 in FIG. 3a). As a result, the spectrum of a single side band superpositioning the lower side band and the upper side band is shown(305 in FIG. 3a), and the bandwidth of the first intermediate frequency signal is reduced to half of a spectrum bandwidth of the high frequency signal received by antenna 101. Therefore, the baseband signals generated on lines 112 and 120, (i.e., the outputs of second down-mixers 111 and 119) are applied to low pass filters 113 and 121 as channel filters which pass, with minimal attenuation, frequency components of the baseband signal up to the low pass filter cutoff frequency, and reject frequency components of the baseband signal above the low pass filter cutoff frequency. Low pass filters 113 and 121 generate filtered signals of the bandwidth of the single side band on lines 114 and 122 which are coupled to blocking capacitors 115 and 123, respectively. Blocking capacitors 115 and 123 are operative to block the DC components of the baseband level signals filtered by low pass filters 113 and 121. The AC components of the baseband level signals generated on lines 116 and 124 are applied to up-mixer 117 of an I-channel and up-mixer 125 of a Q-channel, respectively.
An oscillating signal generated by third local oscillator 139 is applied to 90.degree. phase shifter 140 which generates an oscillating signal with a phase delay of 0.degree. on line 142 and generates an oscillating signal with a phase delay of 90.degree. on line 141. Mixers 117 and 125 are also supplied with oscillating signals on lines 142 and 141, respectively. The oscillating signals generated on lines 142 and 141 are of similar frequencies, but are offset in phase by ninety degrees. Mixers 117 and 125 generate up-converted signals on lines 118 and 126, respectively which have a phase difference of 90.degree. from one another, center at frequencies of the oscillating signal from third local oscillator 139. The up-converted signals generated on lines 118 and 126 are combined to form the third intermediate frequency signal of a double sided band with a spectrum equal to the high frequency signal received by antenna 101. The third intermediate frequency signal is applied to bandpass filter 127 which generates a filtered signal on line 128 which is applied to limiter 129. The limiter 129 amplifies the amplitude of the filtered, third intermediate frequency signal to the amplitude required for demodulation, and then, applies the amplified third intermediate frequency signal by way of line 130 to demodulator 131. Demodulator 131 demodulates frequency-modulated components of the amplified third intermediate frequency signal on line 130 to a voltage increasing/decreasing signal. The demodulated signal is supplied to audio processor 133. Audio processor 133 allows an original aural signal to be outputted through a transducer such as speaker 135 by performing functions, such as, for example, audio filtering, de-emphasis, and variable gain for the demodulated signal.
As is apparent from the foregoing, the Barber receiver shown in FIG. 2 is about 1.5 times as complex as the heterodyne receiver shown in FIG. 1. The heterodyne receiver of type shown in FIG. 1, has other disadvantages. For example, since bandpass filter 161, as a channel filter, cannot be integrated to a semiconductor circuit, because it requires a relatively large amount of space. As such, it suffers from a loss of the passband and distortion of high frequency components to obtain desired suppression properties of an adjacent channel. This results in relatively poor performance of the receiver.
The Barber receiver, shown in FIG. 2, has been designed to resolve the above-mentioned problems with the heterodyne receiver shown in FIG. 1. Namely, low pass filters 113 and 121 in FIG. 2 can be easily embodied by a resistor, a capacitor and an operational amplifier, etc. without using an inductor, so that low pass filters 113 and 121 can be integrated to a semiconductor circuit. Therefore, when a Barber receiver is employed, the occupying space of a hardware is advantageously and relatively reduced in comparison with heterodyne receiver, and loss of a passband and distortion of high frequency components are relatively lessened in comparison with the same suppression properties of an adjacent channel as required in the heterodyne receiver. Notwithstanding the advantages of the Barber receiver, blocking capacitors 115 and 123 are disposed between low pass filters 113, 121 and up-mixers 117, 125, respectively in order to block DC error voltage of output from down-mixers 111 and 119 due to undesired voltage or interference, and DC voltage offset due to bias within a circuit, when the first intermediate frequency signal is generated by using the oscillating signal from second local oscillator 146 having the same frequency as that of the first intermediate frequency signal. Since blocking capacitors 115 and 123 cannot help but have a limited capacity value, they have only low frequency blocking properties. That is, if the capacity value of blocking capacitors 115 and 123 is exceedingly large, it is difficult to integrate blocking capacitors 115 and 123 into a semiconductor circuit and the seizing speed of an initial signal is also very slow.
A frequency response graph shown in FIG. 3c shows the combined frequency response 309 of the low frequency properties(308 in FIG. 3c) and the low pass restricting properties(307 in FIG. 3b).
In FIG. 2, the spectrum (311 in FIG. 3c) of a third intermediate frequency signal, which is up-converted at up-mixers 117 and 125, and then applied through lines 118 and 126 to band pass filter 127, is extended to a low side band and an upper side band from side to side, with the center frequency of a single side band spectrum (309 in FIG. 3c) of a baseband level being translated to a frequency (310 in FIG. 3c) of the oscillating signal from third local oscillator 139 at 0 Hz. The spectrum of a third intermediate frequency signal also represents a distorted shape having a middle depressed portion (312 in FIG. 3c) with increased attenuation toward the center frequency. Therefore, in a frequency-modulated system in which frequency deviation becomes large according to a low amplitude of low frequency components close to DC components at a I-channel and a Q-channel, (i.e., an original aural signal), there have been problems in that the smaller the components whose frequency deviation is small gets, the larger the attenuation the components have, thereby exceeding acceptable distortion levels due to the DC error voltage.
An approach to resolve this problem is more specifically described in U.S. Pat. No. 5,428,836 issued Jun. 27, 1995 and entitled "RADIO RECEIVER FOR FORMING A BASEBAND SIGNAL OF TIME-VARYING FREQUENCIES"(Motorola Inc.). FIG. 4 is a block diagram showing the inner construction of a receiver including the construction shown in FIG. 8 of U.S. Pat. No. 5,428,836. The receiver shown in U.S. Pat. No. 5,428,836 is characterized in that modulator 155 is disposed between first down-mixer 105 and first local oscillator 152 of the existing elements of the Barber receiver shown in FIG. 2.
Turning now to the block diagram of FIG. 4, modulator 155 receives any constant AC signal on line 154 and modulates the oscillating frequency signal (fc in FIG. 5) from first local oscillator 152 by the AC signal on line 154, and then supplies a time-varying frequency signal (503 in FIG. 5) in which the frequency is dependent upon time to first down-mixer 105. Therefore, the second intermediate frequency signal at 0 Hz cannot regularly drop in a low frequency blocking area (308 in FIG. 3c), and the third intermediate frequency signal cannot regularly drop in a middle depressed portion (312 in FIG. 3c). As a result, the aural signal restored and outputted through demodulator 131 and audio processor 133, and then, by way of speaker 135 cannot be regularly attenuated, which is a characteristic of the receiver shown in U.S. Pat. No. 5,428,836.
However, in a frequency-modulated communication system in which the ratio of a signal to noise as disclosed in U.S. Pat. No. 5,428,836 is proportional to the degree of frequency deviation, the time-varying frequency signal on line 156, modulated by the AC signal on line 154, allows the degree of frequency deviation of the first intermediate frequency signal on line 106 to be increased or decreased in a period (502 in FIG. 5) by an amplitude (501 in FIG. 5) of the modulated signal on line 156. Consequently, since the frequency of the modulated signal on line 156 is carried on the aural signal restored through speaker 135 with the volume of sound proportional to the amplitude (501 in FIG. 5), a compensation circuit for suppressing attenuation properties of a low frequency signal is indispensable to the receiver. Particularly, when the transmitter and receiver together share first local oscillator 152 or other local oscillators, a compensation circuit for suppressing attenuation of a low frequency signal should be inserted at the transmission path, or there is a restriction of having to select a modulation method which doesn't affect the transmitting path. Furthermore, in order to generate a low frequency modulation, a signal source for the low frequency modulation and a modulation circuit should be added.